1
LTC3801/LTC3801B
3801f
Micropower
Constant Frequency Step-Down
DC/DC Controllers in ThinSOT
s
High Efficiency: Up to 94%
s
Very Low No-Load Quiescent Current:
Only 16
µ
A (LTC3801)
s
High Output Currents Easily Achieved
s
Internal Soft-Start
s
Wide V
IN
Range: 2.4V to 9.8V
s
Low Dropout: 100% Duty Cycle
s
Constant Frequency 550kHz Operation
s
Burst Mode
®
Operation for High Efficiency
at Light Loads (LTC3801)
s
Burst Mode Operation Disabled for Lower Output
Ripple at Light Loads (LTC3801B)
s
Output Voltage as Low as 0.8V
s
±
1.5% Voltage Reference Accuracy
s
Current Mode Operation for Excellent Line and Load
Transient Response
s
Only 6
µ
A Supply Current in Shutdown (LTC3801)
s
Low Profile (1mm) SOT-23 Package
s
1- or 2-Cell Li-Ion Battery-Powered Applications
s
Wireless Devices
s
Portable Computers
s
Distributed Power Systems
, LTC and LT are registered trademarks of Linear Technology Corporation.
550kHz Micropower Step-Down DC/DC Controller
The LTC
®
3801/LTC3801B are constant frequency cur-
rent mode step-down DC/DC controllers in a low profile
(1mm max) 6-lead SOT-23 (ThinSOT
TM
) package. The
parts provide excellent AC and DC load and line regula-
tion with
±
1.5% output voltage accuracy. The LTC3801
consumes only 195
µ
A of quiescent current in normal
operation, dropping to 16
µ
A under no-load conditions.
The LTC3801/LTC3801B incorporate an undervoltage lock-
out feature that shuts down the device when the input
voltage falls below 2.2V. The LTC3801 automatically
switches into Burst Mode operation at light loads which
enhances efficiency at low output current. In the LTC3801B,
Burst Mode operation is disabled for lower output ripple at
light loads.
To further maximize the life of a battery source, the
external P-channel MOSFET is turned on continuously in
dropout (100% duty cycle). High switching frequency of
550kHz allows the use of a small inductor.
Burst Mode is a registered trademark of Linear Technology Corporation.
ThinSOT is a trademark of Linear Technology Corporation.
I
TH
/RUN
LTC3801/
LTC3801B
10k
402k
866k
0.025
4.7
µ
H
220pF
GND
V
FB
3801 TA01
PGATE
V
IN
SENSE
10
µ
F
V
IN
2.7V TO 9.8V
V
OUT
2.5V
2A
47
µ
F
+
LTC3801 Efficiency vs Load Current*
LOAD CURRENT (mA)
0.1
70
EFFICIENCY (%)
75
80
85
90
1
10
100
10000
1000
3801 TA02
65
60
55
50
95
100
V
IN
= 3.3V
V
IN
= 6.6V
V
IN
= 8.4V
V
IN
= 9.8V
V
OUT
= 2.5V
V
IN
= 4.2V
FEATURES
DESCRIPTIO
U
APPLICATIO S
U
TYPICAL APPLICATIO
U
*SEE NO-LOAD I
Q
vs INPUT VOLTAGE ON THE LAST PAGE OF THIS DATA SHEET
2
LTC3801/LTC3801B
3801f
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Input Voltage Range
q
2.4
9.8
V
Input DC Supply Current
Typicals at V
IN
= 4.2V (Note 4)
Normal Operation
2.4V
V
IN
9.8V, V
ITH
/RUN = 1.3V
195
300
µ
A
SLEEP Mode
2.4V
V
IN
9.8V (LTC3801 Only)
16
30
µ
A
Shutdown
2.4V
V
IN
9.8V, V
ITH
/RUN = 0V (LTC3801)
6
15
µ
A
2.4V
V
IN
9.8V, V
ITH
/RUN = 0V (LTC3801B)
8
17
µ
A
UVLO
V
IN
< UVLO Threshold
1
2
µ
A
Undervoltage Lockout Threshold
V
IN
Rising
q
1.8
2.3
V
V
IN
Falling
q
1.7
2.2
V
Start-Up Current Source
V
ITH
/RUN = 0V (LTC3801)
0.5
1
1.5
µ
A
V
ITH
/RUN = 0V (LTC3801B)
1.0
2
3.0
µ
A
Shutdown Threshold (at I
TH
/RUN)
V
ITH
/RUN Rising
q
0.3
0.6
0.95
V
Regulated Feedback Voltage
0
°
C
T
A
85
°
C (Note 5)
0.788
0.800
0.812
V
40
°
C
T
A
85
°
C (Note 5)
q
0.780
0.800
0.812
V
Feedback Voltage Line Regulation
2.4V
V
IN
9.8V (Note 5)
0.05
mV/V
Feedback Voltage Load Regulation
I
TH
/RUN Sinking 5
µ
A (Note 5)
2
mV/
µ
A
I
TH
/RUN Sourcing 5
µ
A (Note 5)
2
mV/
µ
A
V
FB
Input Current
(Note 5)
2
10
nA
Overvoltage Protect Threshold
Measured at V
FB
0.850
0.880
0.910
V
Overvoltage Protect Hysteresis
40
mV
Oscillator Frequency
Normal Operation
V
FB
= 0.8V
500
550
650
kHz
Output Short Circuit
V
FB
= 0V
210
kHz
Gate Drive Rise Time
C
LOAD
= 3000pF
40
ns
Gate Drive Fall Time
C
LOAD
= 3000pF
40
ns
Peak Current Sense Voltage
Duty Cycle < 40% (Note 6)
LTC3801
q
109
117
125
mV
LTC3801B
q
95
104
113
mV
Peak Current Sense Voltage in Burst Mode Operation
LTC3801 Only
26
mV
Default Soft-Start Time
0.6
ms
ORDER PART
NUMBER
(Note 1)
Input Supply Voltage (V
IN
)........................ 0.3V to 10V
SENSE
, PGATE Voltages ............ 0.3V to (V
IN
+ 0.3V)
V
FB
, I
TH
/RUN Voltages ............................. 0.3V to 2.4V
PGATE Peak Output Current (<10
µ
s) ........................ 1A
Operating Temperature Range (Note 2) .. 40
°
C to 85
°
C
Junction Temperature (Note 3) ............................ 150
°
C
Storage Temperature Range ................. 65
°
C to 150
°
C
Lead Temperature (Soldering, 10 sec).................. 300
°
C
LTC3801ES6
LTC3801BES6
T
JMAX
= 150
°
C,
JA
= 230
°
C/W
The
q
indicates specifications which apply over the full operating
temperature range, otherwise specifications are at T
A
= 25
°
C. V
IN
= 4.2V unless otherwise noted. (Note 2)
ABSOLUTE
M
AXI
M
U
M
RATINGS
W
W
W
U
PACKAGE/ORDER I
N
FOR
M
ATIO
N
W
U
U
Consult LTC Marketing for parts specified with wider operating temperature ranges.
I
TH
/RUN 1
GND 2
V
FB
3
6 PGATE
5 V
IN
4 SENSE
TOP VIEW
S6 PACKAGE
6-LEAD PLASTIC TSOT-23
S6 PART MARKING
LTACR
LTAHN
ELECTRICAL CHARACTERISTICS
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC3801ES6/LTC3801BES6 are guaranteed to meet specifica-
tions from 0
°
C to 70
°
C. Specifications over the 40
°
C to 85
°
C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: T
J
is calculated from the ambient temperature T
A
and power
dissipation P
D
according to the following formula:
T
J
= T
A
+ (P
D
·
JA
°
C/W)
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC3801/LTC3801B are tested in a feedback loop that servos
V
FB
to the output of the error amplifier while maintaining I
TH
/RUN at the
midpoint of the current limit range.
Note 6: Peak current sense voltage is reduced dependent on duty cycle as
given in Figure 1.
3
LTC3801/LTC3801B
3801f
TYPICAL PERFOR A CE CHARACTERISTICS
U
W
Input DC Supply Current (Normal)
vs Input Voltage
V
IN
(V)
2
I
IN
(
µ
A)
205
215
225
5
7
10
3801 G01
195
185
175
3
4
6
8
9
T
A
= 25
°
C
V
ITH
/RUN = 1.3V
V
IN
(V)
2
I
IN
(
µ
A)
16
18
20
5
7
10
3801 G02
14
12
10
3
4
6
8
9
T
A
= 25
°
C
V
IN
(V)
2
I
IN
(
µ
A)
9
12
15
5
7
10
3801 G03
6
3
0
3
4
6
8
9
T
A
= 25
°
C
V
ITH
/RUN = 0V
LTC3801B
LTC3801
Input DC Supply Current (SLEEP)
vs Input Voltage (LTC3801 Only)
Input DC Supply Current
(Shutdown) vs Input Voltage
Undervoltage Lockout Threshold
vs Temperature
Shutdown Threshold
vs Temperature
Regulated Feedback Voltage
vs Temperature
TEMPERATURE (
°
C)
50
30
1.2
V
IN
(V)
1.6
2.2
10
30
50
3801 G04
1.4
2.0
1.8
10
70
90
V
IN
RISING
V
IN
FALLING
TEMPERATURE (
°
C)
50
400
V
ITH
/RUN (mV)
500
600
700
800
30
10
10
30
3801 G05
50
70
90
V
IN
= 4.2V
TEMPERATURE (
°
C)
50
V
FB
(mV)
804
808
812
10
50
3801 G06
800
796
30
10
30
80
90
792
788
V
IN
= 4.2V
Regulated Feedback Voltage
vs Input Voltage
Oscillator Frequency
vs Temperature
Oscillator Frequency
vs Input Voltage
V
IN
(V)
2
788
V
FB
(mV)
792
796
800
804
4
6
8
10
3801 G07
808
812
3
5
7
9
T
A
= 25
°
C
TEMPERATURE (
°
C)
50
500
f
OSC
(kHz)
510
530
540
550
600
570
10
30
50
3801 G08
520
580
590
560
30
10
70
90
V
IN
= 4.2V
V
IN
(V)
2
540
f
OSC
(kHz)
545
550
555
560
3
4
5
6
3801 G09
7
8
9
10
T
A
= 25
°
C
4
LTC3801/LTC3801B
3801f
U
U
U
PI FU CTIO S
I
TH
/RUN (Pin 1): This pin performs two functions. It
serves as the error amplifier compensation point as well as
the run control input. Nominal voltage range for this pin is
0.7V to 1.9V. Forcing this pin below 0.6V causes the
device to be shut down. In shutdown, all functions are
disabled and the PGATE pin is held high.
GND (Pin 2): Ground Pin.
V
FB
(Pin 3): Receives the feedback voltage from an exter-
nal resistor divider across the output.
SENSE
(Pin 4): Current Sense Pin. An external sense
resistor is connected between this pin and V
IN
(Pin 5).
V
IN
(Pin 5): Supply Pin. This pin must be closely de-
coupled to GND (Pin 2).
PGATE (Pin 6): Gate Drive for the External P-Channel
MOSFET. This pin swings from 0V to V
IN
.
+
+
0.15V
SLEEP
COMPARATOR
1.2V
BURST
CLAMP
BURST
DEFEAT
(LTC3801B)
SHUTDOWN
COMPARATOR
ERROR
AMPLIFIER
OVERVOLTAGE
COMPARATOR
SHORT-CIRCUIT
DETECT
SLEEP
SHDN
UV
1
µ
A (LTC3801)
2
µ
A (LTC3801B)
I
TH
/RUN
0.225V
0.8V
0.88V
0V
PGATE
+
SOFT-START
CLAMP
550kHz
OSCILLATOR
SLOPE
COMPENSATION
CURRENT
COMPARATOR
UNDERVOLTAGE
LOCKOUT
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
0.3V
+
+
+
1
V
IN
5
+
VOLTAGE
REFERENCE
0.8V
I
TH
BUFFER
0.3V
R
S
RS
LATCH
V
IN
FREQUENCY
FOLDBACK
Q
4
6
V
FB
3801 FD
GND
3
2
SENSE
15mV (LTC3801B)
I
LIM
BURST
DEFEAT
(LTC3801B)
FU CTIO AL DIAGRA
U
U
W
5
LTC3801/LTC3801B
3801f
OPERATIO
U
(Refer to the Functional Diagram)
Main Control Loop (Normal Operation)
The LTC3801/LTC3801B are constant frequency current
mode step-down switching regulator controllers. During
normal operation, an external P-channel MOSFET is turned
on each cycle when the oscillator sets the RS latch and
turned off when the current comparator resets the latch.
The peak inductor current at which the current comparator
trips is controlled by the voltage on the I
TH
/RUN pin, which
is the output of the error amplifier. The negative input to
the error amplifier is the output feedback voltage V
FB
which is generated by an external resistor divider con-
nected between V
OUT
and ground. When the load current
increases, it causes a slight decrease in V
FB
relative to the
0.8V reference, which in turn causes the I
TH
/RUN voltage
to increase until the average inductor current matches the
new load current.
The main control loop is shut down by pulling the I
TH
/RUN
pin to ground. Releasing the I
TH
/RUN pin allows an
internal 1
µ
A current source (2
µ
A on LTC3801B) to charge
up the external compensation network. When the I
TH
/
RUN pin voltage reaches approximately 0.6V, the main
control loop is enabled and the I
TH
/RUN voltage is pulled
up by a clamp to its zero current level of approximately
one diode voltage drop (0.7V). As the external compensa-
tion network continues to charge up, the corresponding
peak inductor current level follows, allowing normal op-
eration. The maximum peak inductor current attainable is
set by a clamp on the I
TH
/RUN pin at 1.2V above the zero
current level (approximately 1.9V).
Burst Mode Operation (LTC3801 Only)
The LTC3801 incorporates Burst Mode operation at low
load currents (<25% of I
MAX
). In this mode, an internal
clamp sets the peak current of the inductor at a level cor-
responding to an I
TH
/RUN voltage 0.3V above its zero
current level (approximately 1V), even though the actual
I
TH
/RUN voltage is lower. When the inductor's average
current is greater than the load requirement, the voltage at
the I
TH
/RUN pin will drop. When the I
TH
/RUN voltage falls
to 0.15V above its zero current level (approximately 0.85V),
the sleep comparator will trip, turning off the external
MOSFET. In sleep, the input DC supply current to the IC is
reduced to 16
µ
A from 195
µ
A in normal operation. With the
switch held off, average inductor current will decay to zero
and the load will eventually cause the error amplifier out-
put to start drifting higher. When the error amplifier output
rises to 0.225V above its zero current level (approximately
0.925V), the sleep comparator will untrip and normal op-
eration will resume. The next oscillator cycle will turn the
external MOSFET on and the switching cycle will repeat.
Low Load Current Operation (LTC3801B Only)
Under very light load current conditions, the I
TH
/RUN pin
voltage will be very close to the zero current level of 0.85V.
As the load current decreases further, an internal offset at
the current comparator input will ensure that the current
comparator remains tripped (even at zero load current)
and the regulator will start to skip cycles, as it must, in
order to maintain regulation. This behavior allows the
regulator to maintain constant frequency down to very
light loads, resulting in less low frequency noise genera-
tion over a wide load current range.
Figure 1 illustrates this result for the circuit on the front
page of this data sheet using both an LTC3801 (in Burst
Mode operation) and an LTC3801B (with Burst Mode
operation disabled). At an output current of 100mA, the
LTC3801 exhibits an output ripple of 81.6mV
P-P
, whereas
the LTC3801B has an output ripple of only 17.6mV
P-P
. At
lower output current levels, the improvement is even
greater. This comes at a tradeoff of lower efficiency for the
non Burst Mode part at light load currents (see Figure 2).
Also notice the constant frequency operation of the
LTC3801B, even at 5% of maximum output current.
Dropout Operation
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the on cycle decreases. This reduction means that
at some input-output differential, the external P-channel
MOSFET will remain on for more than one oscillator cycle
(start dropping off-cycles) since the inductor current has
not ramped up to the threshold set by the error amplifier.
Further reduction in input supply voltage will eventually
cause the external P-channel MOSFET to be turned on
100%, i.e., DC. The output voltage will then be determined
by the input voltage minus the voltage drop across the
sense resistor, the MOSFET and the inductor.
6
LTC3801/LTC3801B
3801f
V
IN
= 4.2V
5
µ
s/DIV
3801 F01a
V
OUT
= 2.5V
I
OUT
= 100mA
OPERATIO
U
(Refer to the Functional Diagram)
Undervoltage Lockout Protection
To prevent operation of the external P-channel MOSFET
with insufficient gate drive, an undervoltage lockout cir-
cuit is incorporated into the LTC3801/LTC3801B. When
the input supply voltage drops below approximately 1.7V,
the P-channel MOSFET and all internal circuitry other than
the undervoltage block itself are turned off. Input supply
current in undervoltage is approximately 1
µ
A.
Short-Circuit Protection
If the output is shorted to ground, the frequency of the
oscillator is folded back from 550kHz to approximately
210kHz while maintaining the same minimum on time.
V
OUT
Ripple for Front Page Circuit Using the LTC3801
(with Burst Mode Operation)
V
OUT
Ripple for Front Page Circuit Using the LTC3801B
(Burst Mode Operation Disabled)
20mV
AC
/DIV
V
IN
= 4.2V
5
µ
s/DIV
3801 F01b
V
OUT
= 2.5V
I
OUT
= 100mA
20mV
AC
/DIV
Figure 1. Output Ripple Waveforms for the Front Page Circuit
LOAD CURRENT (mA)
0.1
10
1
70
EFFICIENCY (%)
75
80
85
90
100
1000
10000
3801 F02
65
60
55
50
95
100
V
OUT
= 2.5V
V
IN
= 3.3V
V
IN
= 4.2V
V
IN
= 6.6V
V
IN
= 9.8V
V
IN
= 8.4V
Figure 2. LTC3801B Efficiency vs Load Current
This lower frequency allows the inductor current to safely
discharge, thereby preventing current runaway. After the
short is removed, the oscillator frequency will gradually
increase back to 550kHz as V
FB
rises through 0.3V on its
way back to 0.8V.
Overvoltage Protection
If V
FB
exceeds its regulation point of 0.8V by more than
10% for any reason, such as an output short circuit to a
higher voltage, the overvoltage comparator will hold the
external P-channel MOSFET off. This comparator has a
typical hysteresis of 40mV.
Slope Compensation and Inductor's Peak Current
The switch on duty cycle in normal operation is given by:
Duty
V
V
V
D
IN
D
Cycle =
V
OUT
+
+
where V
D
is the forward voltage drop of the external diode
at the average inductor current. For duty cycles less than
40%, the inductor's peak current is determined by:
I
V
V
R
MAX
ITH RUN
SENSE
=
/
.
0 7
10
However, for duty cycles greater than 40%, slope com-
pensation begins and effectively reduces the peak
7
LTC3801/LTC3801B
3801f
OPERATIO
U
(Refer to the Functional Diagram)
inductor current. The amount of reduction is given by the
curve in Figure 3.
Soft-Start
An internal default soft-start circuit is employed at power-
up and/or when coming out of shutdown. The soft-start
circuit works by internally clamping the voltage at the
I
TH
/RUN pin to the corresponding zero current level and
gradually raising the clamp voltage such that the minimum
time required for the programmed switch current to reach
its maximum is approximately 0.6ms. After the soft-start
circuit has timed out, it is disabled until the part is put in
shutdown again or the input supply is cycled.
Figure 3. Maximum Current Limit Trip Voltage vs Duty Cycle
DUTY CYCLE (%)
20
TRIP VOLTAGE (mV)
75
95
100
3801 F03
55
35
40
60
80
30
50
70
90
115
65
85
45
105
LTC3801 SLOPE FACTOR (%)
70
50
30
100
60
80
40
90
V
IN
= 4.2V
T
A
= 25
°
C
LTC3801
LTC3801B
APPLICATIO S I FOR ATIO
W
U
U
U
The basic LTC3801/LTC3801B application circuit is shown
on the front page of this data sheet. External component
selection is driven by the load requirement and begins with
the selection of the inductor and R
SENSE
. Next, the power
MOSFET and the output diode are selected followed by the
input bypass capacitor C
IN
and output bypass capacitor
C
OUT
.
R
SENSE
Selection for Output Current
R
SENSE
is chosen based on the required output current.
With the current comparator monitoring the voltage
developed across R
SENSE
, the threshold of the compara-
tor determines the inductor's peak current. The output
current the LTC3801 can provide is given by:
I
R
I
OUT
SENSE
RIPPLE
=
-
0 117
2
.
where I
RIPPLE
is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section). For the LTC3801B
use 104mV in the previous equation and follow through
the analysis using that number.
A reasonable starting point for setting ripple current is
I
RIPPLE
= (0.4)(I
OUT
). Rearranging the above equation, it
becomes:
R
I
SENSE
OUT
=
<
1
10
(
)(
)
for Duty Cycle
40%
However, for operation that is above 40% duty cycle, slope
compensation effect has to be taken into consideration to
select the appropriate value to provide the required amount
of current. Using Figure 3, the value of R
SENSE
is:
R
SF
I
SENSE
OUT
=
(
)(
)(
)
10
100
where SF is the "Slope Factor."
Inductor Value Calculation
The operating frequency and inductor selection are inter-
related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripple current. However, this is at the expense of efficiency
due to an increase in MOSFET gate charge losses.
The inductance value also has a direct effect on ripple cur-
rent. The ripple current, I
RIPPLE
, decreases with higher in-
ductance or frequency and increases with higher V
IN
or
V
OUT
. The inductor's peak-to-peak ripple current is given by:
I
V
V
f L
V
V
V
V
RIPPLE
IN
OUT
OUT
D
IN
D
=
-
+
+
( )
where f is the operating frequency. Accepting larger values
of I
RIPPLE
allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is
8
LTC3801/LTC3801B
3801f
APPLICATIO S I FOR ATIO
W
U
U
U
I
RIPPLE
= 0.4(I
OUT(MAX)
). Remember, the maximum I
RIPPLE
occurs at the maximum input voltage.
In Burst Mode operation on the LTC3801, the ripple
current is normally set such that the inductor current is
continuous during the burst periods. Therefore, the peak-
to-peak ripple current must not exceed:
I
R
RIPPLE
SENSE
0 03
.
This implies a minimum inductance of:
L
V
V
f
R
V
V
V
V
MIN
IN
OUT
SENSE
OUT
D
IN
D
=
-
+
+
0 03
.
(Use V
IN(MAX)
= V
IN
)
A smaller value than L
MIN
could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot af-
ford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool M
µ
®
cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will in-
crease. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core material saturates "hard," which means that
inductance collapses abruptly when the peak design cur-
rent is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive
than ferrite. A reasonable compromise from the same
Kool M
µ
is a registered trademark of Magnetics, Inc.
manufacturer is Kool M
µ
. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, new designs for surface mount that do
not increase the height significantly are available.
Power MOSFET Selection
An external P-channel power MOSFET must be selected
for use with the LTC3801/LTC3801B. The main selection
criteria for the power MOSFET are the threshold voltage
V
GS(TH)
and the "on" resistance R
DS(ON)
, reverse transfer
capacitance C
RSS
and total gate charge.
Since the LTC3801/LTC3801B are designed for operation
down to low input voltages, a sublogic level threshold
MOSFET (R
DS(ON)
guaranteed at V
GS
= 2.5V) is required
for applications that work close to this voltage. When
these MOSFETs are used, make sure that the input supply
to the LTC3801/LTC3801B is less than the absolute maxi-
mum V
GS
rating, typically 8V.
The required minimum R
DS(ON)
of the MOSFET is governed
by its allowable power dissipation. For applications that may
operate the LTC3801/LTC3801B in dropout, i.e., 100% duty
cycle, at its worst case the required R
DS(ON)
is given by:
R
P
I
p
DS ON
P
OUT MAX
DC
(
)
(
)
%
=
=
(
)
+
(
)
100
2
1
where P
P
is the allowable power dissipation and
p is the
temperature dependency of R
DS(ON)
. (1 +
p) is generally
given for a MOSFET in the form of a normalized R
DS(ON)
vs
temperature curve, but
p = 0.005/
°
C can be used as an
approximation for low voltage MOSFETs.
In applications where the maximum duty cycle is less than
100% and the LTC3801/LTC3801B are in continuous
mode, the R
DS(ON)
is governed by:
R
P
DC I
p
DS ON
P
OUT
(
)
( )
+
(
)
2
1
where DC is the maximum operating duty cycle of the
LTC3801/LTC3801B.
9
LTC3801/LTC3801B
3801f
APPLICATIO S I FOR ATIO
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Output Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As V
IN
approaches V
OUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition the diode must
safely handle I
PEAK
at close to 100% duty cycle. Therefore,
it is important to adequately specify the diode peak current
and average power dissipation so as not to exceed the
diode ratings.
Under normal load conditions, the average current con-
ducted by the diode is:
I
V
V
V
V
I
D
IN
OUT
IN
D
OUT
=
-
+
The allowable forward voltage drop in the diode is calcu-
lated from the maximum short-circuit current as:
V
P
I
F
D
SC MAX
(
)
where P
D
is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to keep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
An additional consideration in applications where low no-
load quiescent current is critical is the reverse leakage
current of the diode at the regulated output voltage. A
leakage greater than several microamperes can represent
a significant percentage of the total input current.
C
IN
and C
OUT
Selection
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (V
OUT
+ V
D
)/
(V
IN
+ V
D
). To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
be used. The maximum RMS capacitor current is given by:
C
I
I
V
V
V
V
IN
RMS
MAX
OUT
IN
OUT
IN
Required
-
(
)
[
]
1 2
/
This formula has a maximum value at V
IN
= 2V
OUT
, where
I
RMS
= I
OUT
/2. This simple worst-case condition is com-
monly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer's
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC3801/LTC3801B, ceramic
capacitors can also be used for C
IN
. Always consult the
manufacturer if there is any question.
The selection of C
OUT
is driven by the required effective
series resistance (ESR). Typically, once the ESR require-
ment is satisfied, the capacitance is adequate for filtering.
The output ripple (
V
OUT
) is approximated by:
V
I
ESR
fC
OUT
RIPPLE
OUT
+
1
8
where f is the operating frequency, C
OUT
is the output
capacitance and I
RIPPLE
is the ripple current in the induc-
tor. The output ripple is highest at maximum input voltage
since
I
L
increases with input voltage.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through-
hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for C
OUT
has been
met, the RMS current rating generally far exceeds the
I
RIPPLE(P-P)
requirement.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec-
trolytic and dry tantalum capacitors are both available in
10
LTC3801/LTC3801B
3801f
APPLICATIO S I FOR ATIO
W
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surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON,
Nichicon PL series and Panasonic SP.
Setting Output Voltage
The LTC3801/LTC3801B develop a 0.8V reference voltage
between the feedback (Pin 3) terminal and ground (see
Figure 4). By selecting resistor R1, a constant current is
caused to flow through R1 and R2 to set the overall output
voltage. The regulated output voltage is determined by:
V
R
R
OUT
=
+
0 8 1
2
1
.
For most applications, an 80k resistor is suggested for R1.
In applications where low no-load quiescent current is
critical, R1 should be made >400k to limit the feedback
divider current to approximately 10% of the chip quiescent
current. If R2 then results in a very high impedance, it may
be beneficial to bypass R2 with a 5pF to 10pF capacitor. To
prevent stray pickup, locate resistors R1 and R2 close to
LTC3801/LTC3801B.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3801/LTC3801B circuits: 1) LTC3801/
LTC3801B DC bias current, 2) MOSFET gate charge cur-
rent, 3) I
2
R losses and 4) voltage drop of the output diode.
1. The V
IN
current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. V
IN
current results in a small loss
which increases with V
IN
.
2. MOSFET gate charge current results from switching the
gate capacitance of the power MOSFET. Each time a
MOSFET gate is switched from low to high to low again,
a packet of charge dQ moves from V
IN
to ground. The
resulting dQ/dt is a current out of V
IN
which is typically
much larger than the DC supply current. In continuous
mode, I
GATECHG
= (f)(dQ).
3. I
2
R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but is
"chopped" between the P-channel MOSFET (in series
with R
SENSE)
and the output diode. The MOSFET R
DS(ON)
plus R
SENSE
multiplied by duty cycle can be summed with
the resistances of L and R
SENSE
to obtain I
2
R losses.
4. The output diode is a major source of power loss at high
currents and gets worse at high input voltages. The
diode loss is calculated by multiplying the forward
voltage times the diode duty cycle multiplied by the load
current. For example, assuming a duty cycle of 50%
with a Schottky diode forward voltage drop of 0.4V, the
loss increases from 0.5% to 8% as the load current
increases from 0.5A to 2A.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2(V
IN
)
2
I
O(MAX)
C
RSS
(f)
Other losses including C
IN
and C
OUT
ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
Figure 4. Setting Output Voltage
3
V
FB
V
OUT
LTC3801/
LTC3801B
R1
3801 F04
R2
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% (
1 +
2 +
3 + ...)
where
1,
2, etc. are the individual losses as a percent-
age of input power.
11
LTC3801/LTC3801B
3801f
APPLICATIO S I FOR ATIO
W
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Foldback Current Limiting
As described in the Output Diode Selection, the worst-
case dissipation occurs with a short-circuited output
when the diode conducts the current limit value almost
continuously. To prevent excessive heating in the diode,
foldback current limiting can be added to reduce the
current in proportion to the severity of the fault.
Foldback current limiting is implemented by adding di-
odes D
FB1
and D
FB2
between the output and the I
TH
/RUN
pin as shown in Figure 5. In a hard short (V
OUT
= 0V), the
current will be reduced to approximately 50% of the
maximum output current.
Figure 5. Foldback Current Limiting
V
FB
I
TH
/RUN
V
OUT
LTC3801/
LTC3801B
R1
3801 F05
R2
D
FB1
D
FB2
+
U
PACKAGE DESCRIPTIO
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
1.50 1.75
(NOTE 4)
2.80 BSC
0.30 0.45
6 PLCS (NOTE 3)
DATUM `A'
0.09 0.20
(NOTE 3)
S6 TSOT-23 0302
2.90 BSC
(NOTE 4)
0.95 BSC
1.90 BSC
0.80 0.90
1.00 MAX
0.01 0.10
0.20 BSC
0.30 0.50 REF
PIN ONE ID
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
3.85 MAX
0.62
MAX
0.95
REF
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
1.4 MIN
2.62 REF
1.22 REF
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
12
LTC3801/LTC3801B
3801f
©
LINEAR TECHNOLOGY CORPORATION 2003
LT/TP 1103 1K · PRINTED IN THE USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
q
FAX: (408) 434-0507
q
www.linear.com
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TYPICAL APPLICATIO
550kHz Micropower Step-Down DC/DC Controller
I
TH
/RUN
LTC3801/
LTC3801B
10k
402k
866k
0.025
4.7
µ
H
220pF
GND
V
FB
3801 TA01
PGATE
V
IN
SENSE
10
µ
F
V
IN
2.7V TO 9.8V
V
OUT
2.5V
2A
47
µ
F
+
V
IN
INPUT VOLTAGE (V)
3
4
15
V
IN
SUPPLY CURRENT (
µ
A)
19
25
5
7
8
3801 TA04
17
23
21
6
9
10
V
OUT
= 2.5V
FRONT PAGE APPLICATION
LTC3801 No-Load I
Q
vs Input Voltage*
*SEE THE FRONT PAGE OF THIS DATA SHEET FOR THE EFFICIENCY vs LOAD CURRENT CURVE
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