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FUNCTIONAL BLOCK DIAGRAM
OUTPUT SIDE
CONTROL
CIRCUIT
INPUT SIDE
CONTROL
CIRCUIT
EMI FILTER
OUTPUT
FILTER
SENSE
REF
+ SENSE
ADJUST
STATUS
V
AUX
INHIBIT
SYNC
I
SHARE
TEMP
V
IN
+V
IN
FIXED
FREQUENCY
DUAL
INTERLEAVED
POWER TRAIN
SENSE
RETURN
RETURN
+V
OUT
+V
OUT
SENSE
REF
ADDC02828SA
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
a
ADDC02828SA
FEATURES
28 V dc Input, 28 V dc @ 3.6 A, 100 W Output
Integral EMI Filter Designed to Meet MIL-STD-461D
Low Weight: 80 Grams
NAVMAT Derated
Many Protection and System Features
APPLICATIONS
Commercial and Military Airborne Electronics
Missile Electronics
Space-Based Antennae and Vehicles
Mobile/Portable Ground Equipment
Distributed Power Architecture for Active Array Radar
GENERAL DESCRIPTION
The ADDC02828SA hybrid dc/dc converter with integral EMI
filter offers the highest power density of any dc/dc converter
available today with its features and in its power range. The
converter with integral EMI filter is a fixed frequency, 1 MHz,
square wave switching dc/dc power supply. It is not a variable
frequency resonant converter. In addition to many protection
features, this converter has system level features that allow it to
be used as a component in larger systems as well as a stand-
alone power supply. The unit is designed for high reliability and
high performance applications where saving space and/or weight
is critical.
The ADDC02828SA is available in three screening grades; all
grades use a hermetically sealed, molybdenum based hybrid
package. Contact factory for MIL-STD-883 device availability.
28 V/100 W DC/DC Converter
with Integral EMI Filter
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700
World Wide Web Site: http://www.analog.com
Fax: 617/326-8703
© Analog Devices, Inc., 1997
PRODUCT HIGHLIGHTS
1. 60 W/cubic inch power density with an integral EMI filter
designed to meet all applicable requirements in MIL-STD-
461D when installed in a typical system setup.
2. Light weight: 80 grams
3. Operational and survivable over a wide range of input condi-
tions: 16 V50 V dc; survives low line, high line and positive
and negative transients. See Input Voltage Range section.
4. High reliability; NAVMAT derated
5. Protection Features Include:
Output Overvoltage Protection
Output Short Circuit Current Protection
Thermal Monitor/Shutdown
Input Overvoltage Shutdown
Input Transient Protection
6. System Level Features Include:
Current Sharing for Parallel Operation
Inhibit Control
Output Status Signal
Synchronization for Multiple Units
Input Referenced Auxiliary Voltage Supply
ADDC02828SASPECIFICATIONS
ELECTRICAL CHARACTERISTICS
Case
Test
ADDC02828SA
Parameter
Temp
Level
Conditions
Min
Typ
Max
Units
INPUT CHARACTERISTICS
Steady State Operating Input Voltage Range
1
Full
VI
I
O
= 0.36 A to 3.6 A
18
28
40
V
Abnormal Operating Input Voltage Range
(Per MIL-STD-704D)
1
Full
VI
I
O
= 0.36 A to 2.9 A
16
50
V
Input Overvoltage Shutdown
+25
°
C
I
50
52.5
55
V
No Load Input Current
+25
°
C
VI
85
100
mA
Disabled Input Current
+25
°
C
VI
1
5
mA
OUTPUT CHARACTERISTICS
2, 3
Output Voltage (V
O
)
+25
°
C
I
I
O
= 0.36 A to 3.6 A, V
IN
= 18 V to 40 V dc
27.44 28.00
28.56 V
Full
VI
I
O
= 0.36 A to 3.6 A, V
IN
= 18 V to 40 V dc
26.88
29.12 V
Full
VI
I
O
= 0.36 A to 2.9 A, V
IN
= 16 V to 50 V dc
26.88
29.12 V
Line Regulation
+25
°
C
VI
I
O
= 3.6 A, V
IN
= 18 V to 40 V dc
10
60
mV
Load Regulation
+25
°
C
VI
V
IN
= 28 V dc, I
O
= 0.36 A to 3.6 A
15
45
mV
Output Ripple/Noise
4
+25
°
C
I
I
O
= 3.6 A, 5 kHz 2 MHz BW
50
100
mV p-p
Output Current (I
O
)
Full
VI
V
IN
= 18 V to 40 V dc
0.36
3.6
A
Output Overvoltage Protection
+25
°
C
V
I
O
= 3.6 A, Open Remote Sense Connection
125
% V
O
Nom
Output Current Limit
+25
°
C
V
V
O
= 90% V
OUT
Nom
130
% I
O
max
Output Short Circuit Current
+25
°
C
I
11
A
ISOLATION CHARACTERISTICS
Isolation Resistance
+25
°
C
I
Input to Output or Any Pin to Case at 500 V dc
100
M
DYNAMIC CHARACTERISTICS
4
Maximum Output Voltage Deviation Due to
Step Change in Load
+25
°
C
I
I
O
= 1.8 A to 3.6 A or 3.6 A to 1.8 A, di/dt = 0.5 A/
µ
s
1.8
V
Response Time Due to Step Change in Load
+25
°
C
I
I
O
= 1.8 A to 3.6 A or 3.6 A to 1.8 A, di/dt = 0.5 A/
µ
s
150
µ
s
Time for V
OUT
to Return within 2% of Final Value
Soft Start Turn-On Time
5
+25
°
C
I
I
O
= 3.6 A, From Inhibit High to Status High
15
20
ms
THERMAL CHARACTERISTICS
Efficiency
+25
°
C
I
I
O
= 2.2 A
81
85
%
Full
VI
I
O
= 2.2 A
80
%
+25
°
C
I
I
O
= 3.6 A
81
85
%
Full
VI
I
O
= 3.6 A
80
%
Hottest Junction Temperature
6
+90
°
C
V
I
O
= 3.6 A
110
°
C
CONTROL CHARACTERISTICS
Clock Frequency
Full
VI
I
O
= 0.36 A
0.85
0.99
MHz
ADJUST (Pin 3) V ADJ
+25
°
C
I
5.5
5.6
5.7
V
STATUS (Pin 4)
V
OH
+25
°
C
I
I
OH
= 400
µ
A
2.4
4.0
V
V
OL
+25
°
C
I
I
OL
= 1 mA
0.15
0.7
V
V
AUX
(Pin 5)
V
O
(nom)
+25
°
C
I
I
AUX
= 5 mA, Load Current = 3.6 A
12.85 13.1
13.35 V
INHIBIT (Pin 6)
V
IL
+25
°
C
I
0.5
V
I
IL
+25
°
C
I
V
IL
= 0.5 V
1.2
mA
V
I
(Open Circuit)
+25
°
C
I
15
V
SYNC (Pin 7)
7
V
IH
+25
°
C
I
4.0
V
I
IH
+25
°
C
I
V
IH
= 7.0 V
150
µ
A
I
SHARE
(Pin 8)
+25
°
C
I
I
O
= 3.6 A
2.70
2.80
2.90
V
TEMP (Pin 9)
+25
°
C
V
3.90
V
NOTES
1
50 V dc upper limit rated for transient condition of up to 50 ms. 16 V dc lower limit rated for continuous operation during emergency condition. Steady state and
abnormal input voltage range require source impedance sufficient to ensure input stability at low line. See sections entitled System Instability Considerations and
Input Voltage Range.
2
Measured at the remote sense points.
3
Unit regulates output voltage to zero load.
4
C
LOAD
= 0.
5
Output is fully loaded into a constant resistive load.
6
Refer to Thermal Characteristics section for more information.
7
Unit has internal pull-down; refer to section entitled Pin 7 (SYNC).
Specifications subject to change without notice.
REV. 0
2
(T
C
= +25 C, V
IN
= 28 V dc 0.5 V dc, unless otherwise noted; full temperature range is
55 C to +90 C; all temperatures are case and T
C
is the temperature measured at the center of the package bottom.)
ADDC02828SA
REV. 0
3
ABSOLUTE MAXIMUM RATINGS*
INHIBIT . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V dc, 0.5 V dc
SYNC . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8.0 V dc, 0.5 V dc
I
SHARE
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 V dc, 0.5 V dc
TEMP . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 V dc, 0.3 V dc
Common-Mode Voltage, Input to Output . . . . . . . . . 500 V dc
Lead Soldering Temp (10 sec) . . . . . . . . . . . . . . . . . . . +300
°
C
Storage Temperature . . . . . . . . . . . . . . . . . . 65
°
C to +150
°
C
Maximum Junction Temperature . . . . . . . . . . . . . . . . . +150
°
C
Maximum Case Operating Temperature . . . . . . . . . . . +125
°
C
*Absolute maximum ratings are limiting values, to be applied individually, and
beyond which the serviceability of the circuit may be impaired. Functional
operability under any of these conditions is not necessarily implied. Exposure of
absolute maximum rating conditions for extended periods of time may affect
device reliability.
ORDERING GUIDE
Operating
Temperature
Device
Range (Case)
Description
ADDC02828SAKV
40
°
C to +85
°
C
Hermetic Package
ADDC02828SATV
55
°
C to +90
°
C
Hermetic Package
ADDC02828SATV/883B*
55
°
C to +125
°
C Hermetic Package
*Contact factory.
EXPLANATION OF TEST LEVELS
Test Level
I
100% production tested.
II
100% production tested at +25
°
C, and sample tested at
specified temperatures.
III
Sample tested only.
IV
Parameter is guaranteed by design and characterization
testing.
V
Parameter is a typical value only.
VI
All devices are 100% production tested at +25
°
C. 100%
production tested at temperature extremes for military
temperature devices; guaranteed by design and charac-
terization testing for industrial devices.
PIN FUNCTION DESCRIPTIONS
Pin
No.
Name
Function
1
SENSE
Feedback loop connection for remote sensing
output voltage. Must always be connected to
output return for proper operation.
2
+SENSE
Feedback loop connection for remote sensing
output voltage. Must always be connected to
+V
OUT
for proper operation.
3
ADJUST
Adjusts output voltage setpoint.
4
STATUS
Indicates output voltage is within
±
5% of
nominal. Active high referenced to SENSE
(Pin 1).
5
V
AUX
Low level dc auxiliary voltage supply refer-
enced to input return (Pin 10).
6
INHIBIT
Power Supply Inhibit. Active low and refer-
enced to input return (Pin 10).
7
SYNC
Clock synchronization input for multiple units;
referenced to input return (Pin 10).
8
I
SHARE
Current share pin that allows paralleled units
to share current typically within
±
5% at full
load; referenced to input return (Pin 10).
9
TEMP
Case temperature indicator and temperature
shutdown override; referenced to input return
(Pin 10).
10
V
IN
Input Return.
11
+V
IN
+28 V Nominal Input Bus.
12
+V
OUT
+28 V dc Output.
13
+V
OUT
+28 V dc Output.
14
SENSE
REF
Output Sense Reference.
15
SENSE
REF
Output Sense Reference.
16
RETURN
Output Return.
17
RETURN
Output Return.
PIN CONFIGURATION
1
11
12
17
TOP
VIEW
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Therefore, proper ESD precautions are recommended to avoid performance degradation or loss
of functionality.
WARNING!
ESD SENSITIVE DEVICE
ADDC02828SATypical Performance Curves
REV. 0
4
OUTPUT POWER Volts
88
10
20
30
40
60
70
80
90
86
84
82
80
78
76
74
72
70
68
50
100
V
O
= 28V
T = +25 C
40V
28V
18V
EFFICIENCY %
Figure 1. Efficiency vs. Line and Load at +25
°
C
T
CASE
°
C
87
86
82
55
75
45
EFFICIENCY %
35 25 15 5
5
15
25
35
45
55
65
85
84
83
85
28V
IN
75W PEAK
Figure 2. Efficiency vs. Case Temperature (
°
C)
(at Nominal V
IN
, 75% Max Load)
OUTPUT POWER Watts
14.0
80
75
85
90
95
100
13.9
13.8
13.7
13.6
13.5
13.4
13.3
13.2
13.1
13.0
INPUT VOLTAGE Volts
Figure 3. Low Line Dropout vs. Load at 90
°
C Case
Temperature
T
CASE
°
C
1.00
0.50
1.00
55
75
45 35 25 15 5
5
15
25
35
45
55
65
0.00
0.50
85
28V
IN
75WATTS
V
OUT
DEVIATION %
Figure 4. Output Voltage vs. Case Temperature (
°
C)
10V
10V
V
OUT
V
INHIBIT
5ms
Figure 5. Output Voltage Transient During Turn-On with
Minimum Load Displaying Soft Start When Supply Is
Enabled
50µs
500mV/Div
Figure 6. Output Voltage Transient Response to
a 1.8 A to 3.6 A Step Change in Load with Zero Load
Capacitance
ADDC02828SA
REV. 0
5
FREQUENCY Hz
1.0
10
.001
.01
0.1
100
10k
100k
1k
|Z
OUT
|
Figure 9. Incremental Output Impedance (Magnitude)
1mV
100µV
VOLTS
2.00 MHz/Div
Figure 10. Output Voltage Ripple Spectrum
10
50k
100
1k
0
10
100
20
30
40
50
60
70
80
90
10k
FREQUENCY Hz
|A
S
| dB
Figure 7. Audio Susceptibility (Magnitude of V
OUT
/V
IN
)
FREQUENCY Hz
10.0
10
0.01
0.1
|Z
IN
|
1.0
100
10k
100k
1k
18Vdc
28Vdc
Figure 8. Incremental Input Impedance (Magnitude)
ADDC02828SATypical EMI Curves and Test Setup
REV. 0
6
+V
IN
V
IN
+V
OUT
RETURN
CASE
LISN
1/4
0.1µF
GROUND PLANE
82nF
82nF
2µF
1
100µF
TWO METERS OF
TWISTED CABLE
NOTE: 100µF CAPACITOR AND 1
RESISTOR PROVIDE STABILIZATION FOR 100µH DIFFERENTIAL SOURCE INDUCTANCE
INTRODUCED BY THE LISNs. REFER TO SECTION ON EMI CONSIDERATIONS FOR MORE INFORMATION.
LISN
Figure 15. Schematic of Test Setup for EMI Measurements
FREQUENCY MHz
130
0.01
0.0001
0.001
70
90
30
EMISSION LEVEL dB µV
110
50
CONDUCTED EMISSIONS CE101
CE1011 4.5 AMPS
Figure 11. Conducted Emissions, MIL-STD-461D, CE101,
+28 V Hot Line 100 W Load
FREQUENCY MHz
130
10
0.01
0.1
70
90
30
EMISSION LEVEL dB µV
110
50
1
CONDUCTED EMISSIONS CE102
LIMIT 28VDC
Figure 12. Conducted Emissions, MIL-STD-461D, CE102,
+28 V Hot Line 100 W Load
FREQUENCY MHz
166
0.1
0.0001
0.001
106
126
66
EMISSION LEVEL dB/pT
146
86
0.01
RE101 MILSTD461D
RE1011
Figure 13. Radiated Emissions, MIL-STD-461D, RE101,
100 W Load
FREQUENCY MHz
90
10
0.01
0.1
30
50
30
EMISSION LEVEL dB µV/m
70
10
1
100
1000
RADIATED EMISSIONS RE102
RE1022
Figure 14. Radiated Emissions, MIL-STD-461D, RE102,
Vertical Polarity, 100 W Load
Note: Figures 1115 were obtained from measurements on the ADDC02805SA, a single 5 V dc output converter. Since the con-
struction and topology of the ADDC02828SA is almost identical to the ADDC02805SA, and the component values of the EMI
differential and common-mode filter in the ADDC02828SA are identical to the ADDC02805SA, the subject figures are shown here
as typical of the ADDC02828SA.
ADDC02828SA
REV. 0
7
BASIC OPERATION
The ADDC02828SA converter uses a flyback topology with
dual interleaved power trains operating 180
°
out of phase. Each
power train switches at a fixed frequency of 500 kHz, resulting
in a 1 MHz fixed switching frequency as seen at the input and
output of the converter. In a flyback topology, energy is stored
in the inductor during one half portion of the switching cycle
and is then transferred to the output filter during the next half
portion. With two interleaved power trains, energy is transferred
to the output filter during both halves of the switching cycle,
resulting in smaller filters to meet the required ripple.
A five pole differential input EMI filter, along with a common-
mode EMI capacitor and careful attention to layout parasitics, is
designed to meet all applicable requirements in MIL-STD-461D
when installed in a typical system setup. A more detailed dis-
cussion of CE102 and other EMI issues is included in the sec-
tion entitled EMI Considerations.
The converter uses current mode control and employs a high
performance opto-isolator in its feedback path to maintain isola-
tion between input and output. The control circuit is designed
to give a nearly constant output current as the output voltage
drops from V
O
nom to V
SC
during a short circuit condition. It
does not let the current fold back below the maximum rated
output current. The output overvoltage protection circuitry,
which is independent from the normal feedback loop, protects
the load against a break in the remote sense leads. Remote
sense connections, which can be made at the load, can adjust
for voltage drops of as much as 0.25 V dc between the converter
and the load, thereby maintaining an accurate voltage level at
the load.
An input overvoltage protection feature shuts down the con-
verter when the input voltage exceeds (nominally) 52.5 V dc.
An internal temperature sensor shuts down the unit and pre-
vents it from becoming too hot if the heat removal system fails.
The temperature sensed is the case temperature and is factory
set to trip at a nominal case temperature of 110
°
C to 115
°
C.
The shut-down temperature setting can be raised externally or
disabled by the user.
Each unit has an INHIBIT pin that can be used to turn off the
converter. This feature can be used to sequence the turn-on of
multiple converters and to reduce input power draw during
extended time in a no load condition.
A SYNC pin, referenced to the input return line (Pin 10), is
available to synchronize multiple units to one switching fre-
quency. This feature is particularly useful in eliminating beat
frequencies that may cause increased output ripple on paralleled
units. A current share pin (I
SHARE
) is available that permits
paralleled units to share current, typically within 5% at full load.
A low level dc auxiliary voltage supply referenced to the input
return line is provided for miscellaneous system use.
PIN CONNECTIONS
Pins 1 and 2 ( SENSE)
Pins 1 and 2 must always be connected for proper operation,
although failure to make these connections will not be catastrophic
to the converter under normal operating conditions. If no load
is present on the converter, failure to make these connections
could result in damage to the device. If the ADDC02828SA is
used to provide a +28 V dc output, Pin 1 must always be con-
nected to the output sense reference (Pins 14 and 15) and Pin 2
must always be connected to +V
OUT
(Pins 12 and 13). If the
ADDC02828SA is used to provide a 28 V dc output, Pin 1
must always be connected to RETURN (Pins 16 and 17) and
Pin 2 must always be connected to the output sense reference
(Pins 14 and 15). The connections to +V
OUT
(for a +28 V dc
output) and RETURN (for a 28 V dc output) can be made at
any one of the output pins of the converter, or remotely at the
load. A remote connection at the load can adjust for voltage
drops of as much as 0.25 V dc between the converter and the
load.
Long remote sense leads can affect converter stability, although
this condition is rare. The impedance of the long power leads
between the converter and the remote sense point could affect
the converter's unity gain crossover frequency and phase mar-
gin. Consult factory if long remote sense leads are to be used.
Pin 3 (ADJUST)
An adjustment pin is provided so the user can change the nomi-
nal output voltage during the prototype stage. Since very low
temperature coefficient resistors are used to set the output volt-
age and maintain tight regulation over temperature, using stan-
dard external resistors to adjust the output voltage will loosen
output regulation over temperature. Furthermore, since the
status trip point is not changed when the output voltage is ad-
justed using external resistors, the status line will no longer trip
at the standard levels of the newly adjusted output voltage. If
necessary, modified standard units can be ordered with the
necessary changes made inside the package at the factory. The
ADJUST function is sensitive to noise, and care should be taken
in the routing of connections.
To make the output voltage higher, place a resistor from ADJUST
(Pin 3) to SENSE (Pin 1). To make the output voltage lower,
place a resistor from ADJUST (Pin 3) to +SENSE (Pin 2).
Figures 16 and 17 show resistor values for a
±
5% change in
output voltage:
OUTPUT VOLTAGE %
8
7
RESISTANCE M
1
99
95
98
97
96
5
4
3
2
6
Figure 16. External Resistor Value for Reducing Output
Voltage
ADDC02828SA
REV. 0
8
OUTPUT VOLTAGE %
5
4
RESISTANCE M
0
101
105
102
103
104
3
2
1
Figure 17. External Resistor Value for Increasing Output
Voltage
With regard to the range that the output voltage can be adjusted
by the user, there are two concerns. As the output voltage is
raised it may become difficult to maintain regulation at full
power and low input voltage. As the output voltage is lowered,
it may become difficult to maintain regulation at minimum
power and high input line.
Pin 4 (STATUS)
Pin 4 is active high referenced to SENSE (Pin 1), indicating
that the output voltage is typically within
±
5%. The pin is
pulled both up and down by internal circuitry. Figures 18 and
19 show the typical source and sink capabilities of the status
output. Refer to the paragraphs describing Pin 3 (ADJUST) for
effect on status trip point.
I
OH
mA
5
4
0.2
0.4
0.6
0.8
1
1.2
1.4
3
2
0
1
V
OH
Volts
Figure 18. Source Capability of Status Output
I
OL
mA
1.0
0.8
0.0
1.0
19.0
4.0
7.0
V
OL
V
10.0
13.0
16.0
0.6
0.4
0.2
Figure 19. Sink Capability of Status Output
Pin 5 (V
AUX
)
Pin 5 is referenced to the input return and provides a semi-
regulated 11 V to 14 V dc voltage supply for miscellaneous
system use. The maximum permissible current draw is 5 mA
and the voltage varies with the output load of the converter as
shown in Figure 20.
CONVERTER OUTPUT CURRENT Amps
14
0
1
2
3
4
13
12
10
11
28V
IN
V
AUX
Volts
Figure 20. V
AUX
vs. Load
Pin 6 (INHIBIT)
Pin 6 is active low and is referenced to the input return of the
converter. Connecting it to the input return will turn the
converter off. For normal operation, the inhibit pin is inter-
nally pulled up to 12 V. Use of an open collector circuit is
recommended.
When Pin 6 is disconnected from input return, the converter
will restart in the soft-start mode. Pin 6 must be kept low for at
least 2 milliseconds to initiate a full soft start. Shorter off times
will result in a partial soft start. Figure 21 shows the input
characteristics of Pin 6.
ADDC02828SA
REV. 0
9
V
IL
V
1.2
1.1
0.7
0.5
2.0
1.0
I
IL
mA
1.5
1.0
0.9
0.8
Figure 21. Input Characteristics of Pin 6 When Pulled Low
Pin 7 (SYNC)
Pin 7 can be used for connecting multiple converters to a mas-
ter clock. This master clock can be either an externally user-
supplied clock or it can be a converter that has been modified
and designated as a master unit. Consult factory for availability
of these devices. Capacitive coupling of the clock signal will
ensure that if the master clock stops working the individual
units will continue to operate at their own internal clock fre-
quency, thereby eliminating a potential single point failure.
Capacitive coupling will also permit a wider duty cycle to be
used. The SYNC pin has an internal pull-down so it is not
necessary to sink any current when driving the pin low. Reference
Figure 28 for a fault tolerant, secondary side powered SYNC
drive circuit.
For user-supplied master clocks with no external circuitry, the
following specifications must be met:
a. Frequency: 1.00 MHz min
b. Duty cycle: 7% min, 14% max
c. High state voltage high level: 4 V min to 7 V max
d. Low state voltage low level: 0 V min to 3.0 V max
Users should note that the SYNC pin is referenced to the input
return of the converter. If the user-supplied master clock is
generated on the output side of the converter, the signal should
be isolated.
Users should be careful about the frequency selected for the
external master clock. Higher switching frequencies will reduce
efficiency and may reduce the amount of output power available at
minimum input line. Consult factory for modified standard switch-
ing frequency to accommodate system clock characteristics.
Pin 8 (I
SHARE
)
Pin 8 allows paralleled converters to share the total load cur-
rent, typically within
±
5% at full load. To use the current share
feature, connect all current share pins to each other and con-
nect the SENSE pins on each of the converters. The current
sharing function is sensitive to the differential voltage between the
input return pins of paralleled converters. The current sharing
function is also sensitive to noise, and care should be taken in
the routing of connections. Refer to Figure 27 for typical appli-
cation circuits using paralleled converters.
Pin 9 (TEMP)
Pin 9 can be used to indicate case temperature or to raise or
disable the temperature at which thermal shutdown occurs.
Typically, 3.90 V corresponds to +25
°
C, with a +13.1 mV/
°
C
change for every 1
°
C rise. The sensor IC (connected from Pin 9
to the input return [Pin 10]) has a 13.1 k
impedance.
The thermal shutdown feature has been set to shut down the
converter when the case temperature is nominally 110
°
C to
115
°
C. To raise the temperature at which shutdown occurs,
connect a resistor with the value shown in Figure 22 from Pin 9
to the input return (Pin 10). To completely disable the tem-
perature shutdown feature, connect a 50 k
resistor from Pin 9
to the input return (Pin 10).
SHUTDOWN CASE TEMPERATURE
°
C
1400
0
120
150
125
130
135
140
145
1200
1000
800
400
200
600
RESISTANCE k
Figure 22. External Resistor Value for Raising
Temperature Shutdown Point
INPUT VOLTAGE RANGE
The steady state operating input voltage range for the converter
is defined as 18 V to 40 V. The abnormal operating input volt-
age range is defined as 16 V to 50 V. In accordance with MIL-
STD-704D, the converter can operate up to 50 V dc input for
transient conditions as long as 50 milliseconds, and it can oper-
ate down to 16 V dc input for continuous operation during
emergency conditions. Figure 3 (typical low line dropout vs.
load) shows that the converter can work continuously down to
and below 16 V dc under reduced load conditions.
The ADDC02828SA can be modified to survive, but not work
through, the upper limit input voltages defined in MIL-STD-704A
(aircraft) and MIL-STD-1275A (military vehicles). MIL-STD-704A
defines an 80 V surge that lasts for 1 second before it falls below
50 V, while MIL-STD-1275A defines a 100 V surge that lasts
for 200 milliseconds before it falls below 50 V. In both cases,
the ADDC02828SA can be modified to operate to specification
up to the 50 V input voltage limit and to shut down and protect
itself during the time the input voltage exceeds 50 V. When the
input voltage falls below 50 V as the surge ends, the converter
will automatically initiate a soft start. In order to survive these
higher input voltage surges; the modified converter, however, will
no longer have input transient protection as described below.
Contact the factory for information on units surviving high input
voltage surges.
Input Voltage Transient Protection: The converter has a
transient voltage suppressor connected across its input leads to
protect the unit against high voltage pulses (both positive and
negative) of short duration. With the power supply connected
in the typical system setup shown in Figure 15, a transient
voltage pulse is created across the converter in the following
manner. A 20
µ
F capacitor is first charged to 400 V. It is then
connected directly across the converter's end of the two meter
power lead cable through a 2
on-state resistance MOSFET.
ADDC02828SA
REV. 0
10
The duration of this connection is 10
µ
s. The pulse is repeated
every second for 30 minutes. This test is repeated with the
connection of the 20
µ
F capacitor reversed to create a negative
pulse on the supply leads. (If continuous reverse voltage protec-
tion is required, a diode can be added externally in series at the
expense of lower efficiency for the power system.)
The converter responds to this input transient voltage test by
shutting down due to its input overvoltage protection feature.
Once the pulse is over, the converter initiates a soft-start, which
is completed before the next pulse. No degradation of converter
performance occurs.
THERMAL CHARACTERISTICS
Junction and Case Temperatures: It is important for the
user to know how hot the hottest semiconductor junctions
within the converter get and to understand the relationship
between junction, case and ambient temperatures. The hottest
semiconductors in the 100 W product line of Analog Devices'
high density power supplies are the switching MOSFETs and
the output rectifiers. There is an area inside the main power
transformers that is hotter than these semiconductors, but it is
within NAVMAT guidelines and well below the Curie tempera-
ture of the ferrite. (The Curie temperature is the point at which
the ferrite begins to lose its magnetic properties.)
Since NAVMAT guidelines require that the maximum junction
temperature be 110
°
C, the power supply manufacturer must
specify the temperature rise above the case for the hottest semi-
conductors so the user can determine what case temperature
is required to meet NAVMAT guidelines. The thermal charac-
teristics section of the specification table states the hottest junction
temperature for maximum output power at a specified case
temperature. The unit can operate to higher case temperatures
than 90
°
C, but 90
°
C is the maximum temperature that permits
NAVMAT guidelines to be met.
Case and Ambient Temperatures: It is the user's responsibility
to properly heat sink the power supply in order to maintain the
appropriate case temperature and, in turn, the maximum junction
temperature. Maintaining the appropriate case temperature is a
function of the ambient temperature and the mechanical heat
removal system. The static relationship of these variables is
established by the following formula:
T
C
=
T
A
+
( P
D
×
R
CA
)
where
T
C
= case temperature measured at the center of the pack-
age bottom,
T
A
= ambient temperature of the air available for cooling,
P
D
= the power, in watts, dissipated in the power supply,
R
CA
= the thermal resistance from the center of the package
to free air, or case to ambient.
The power dissipated in the power supply, P
D
, can be calcu-
lated from the efficiency, , given in the data sheets and the
actual output power, P
O
, in the user's application by the follow-
ing formula:
P
D
=
P
O
1
1
For example, at 80 W of output power and 80% efficiency, the
power dissipated in the power supply is 20 W. If, under these
conditions, the user wants to maintain NAVMAT deratings
(i.e., a case temperature of approximately 90
°
C) with an ambi-
ent temperature of 75
°
C, the required thermal resistance, case
to ambient, can be calculated as
90 = 75 + (20
×
R
CA
) or R
CA
= 0.75
°
C/W
This thermal resistance, case to ambient, will determine what
kind of heat sink and whether convection cooling or forced air
cooling is required to meet the constraints of the system.
SYSTEM INSTABILITY CONSIDERATIONS
In a distributed power supply architecture, a power source
provides power to many "point-of-load" (POL) converters. At
low frequencies, the POL converters appear incrementally as
negative resistance loads. This negative resistance could cause
system instability problems.
Incremental Negative Resistance: A POL converter is de-
signed to hold its output voltage constant no matter how its
input voltage varies. Given a constant load current, the power
drawn from the input bus is therefore also a constant. If the
input voltage increases by some factor, the input current must
decrease by the same factor to keep the power level constant.
In incremental terms, a positive incremental change in the
input voltage results in a negative incremental change in the
input current. The POL converter therefore looks, incremen-
tally, as a negative resistor.
The value of this negative resistor at a particular operating
point, V
IN
, I
IN
, is:
R
N
=
V
IN
I
IN
Note that this resistance is a function of the operating point. At
full load and low input line, the resistance is its smallest, while
at light load and high input line, it is its largest.
Potential System Instability: The preceding analysis assumes
dc voltages and currents. For ac waveforms the incremental
input model for the POL converter must also include the effects
of its input filter and control loop dynamics. When the POL
converter is connected to a power source, modeled as a voltage
source, V
S,
in series with an inductor, L
S
, and some positive
resistor, R
S
, the network of Figure 23 results.
L
P
C
P
|R
N
|
ADI DC/DC CONVERTER
L
S
R
S
V
S
INPUT
TERMINALS
Figure 23. Model of Power Source and POL Converter
Connection
The network shown in Figure 23 is second order and has the
following characteristic equation:
s
2
(L
S
+
L
P
)C
+
s
(L
S
+
L
P
)
|R
N
|
+
R
S
C
P
+
1
=
0
ADDC02828SA
REV. 0
11
For the power delivery to be efficient, it is required that R
S
<< R
N
.
For the system to be stable, however, the following relationship
must hold:
C
P
|R
N
|
>
(L
S
+
L
P
)
R
S
or R
S
>
(L
S
+
L
P
)
C
P
|R
N
|
Notice from this result that if (L
S
+ L
P
) is too large, or if R
S
is
too small, the system might be unstable. This condition would
first be observed at low input line and full load since the abso-
lute value of R
N
is smallest at this operating condition.
If an instability results, and it cannot be corrected by changing
L
S
or R
S
(such as during the MIL-STD-461D tests) due to the
LISN requirement, one possible solution is to place a capacitor
across the input of the POL converter. Another possibility is to
place a small resistor in series with this extra capacitor.
The analysis so far has assumed the source of power was a volt-
age source (e.g., a battery) with some source impedance. In
some cases, this source may be the output of a front-end (FE)
converter. Although each FE converter is different, a model for
a typical one would have an LC output filter driven by a voltage
source whose value was determined by the feedback loop. The
LC filter usually has a high Q, so the compensation of the
feedback loop is chosen to help dampen any oscillations that
result from load transients. In effect, the feedback loop adds
"positive resistance" to the LC network.
When the POL converter is connected to the output of this FE
converter, the POL's "negative resistance" counteracts the
effects of the FE's "positive resistance" offered by the feedback
loop. Depending on the specific details, this might simply mean
that the FE converter's transient response is slightly more oscil-
latory, or it may cause the entire system to be unstable.
For the ADDC02828SA, L
P
is approximately 1
µ
H and C
P
is
approximately 4
µ
F. Figure 8 shows a more accurate depiction
of the input impedance of the converter as a function of fre-
quency. The negative resistance is, itself, a very good incremen-
tal model for the power state of the converter for frequencies
into the several kHz range.
NAVMAT DERATING
NAVMAT is a Navy power supply reliability manual that is
frequently cited by specifiers of power supplies. A key section of
NAVMAT P4855-1A discusses guidelines for derating designs
and their components. The two key derating criteria are voltage
derating and power derating. Voltage derating is done to reduce
the possibility of electrical breakdown, whereas power derating
is done to maintain the component material below a specified
maximum temperature. While power deratings are typically
stated in terms of current limits (e.g., derate to x% of maximum
rating), NAVMAT also specifies a maximum junction tem-
perature of the semiconductor devices in a power supply. The
NAVMAT component deratings applicable to the ADDC02828SA
are as follows:
Resistors
80% voltage derating
50% power derating
Capacitors
50% voltage and ripple voltage derating
70% ripple current derating
Transformers and Inductors
60% continuous voltage and current derating
90% surge voltage and current derating
20
°
C less than rated core temperature
30
°
C below insulation rating for hot spot temperature
25% insulation breakdown voltage derating
40
°
C maximum temperature rise
Transistors
50% power derating
60% forward current (continuous) derating
75% voltage and transient peak voltage derating
110
°
C maximum junction temperature
Diodes (Switching, General Purpose, Rectifiers)
70% current (surge and continuous) derating
65% peak inverse voltage derating
110
°
C maximum junction temperature
Diodes (Zeners)
70% surge current derating
60% continuous current derating
50% power derating
110
°
C maximum junction temperature
Microcircuits (Linears)
70% continuous current derating
75% signal voltage derating
110
°
C maximum junction temperature
The ADDC02828SA can meet all the derating criteria listed
above. However, there are a few areas of the NAVMAT deratings
where meeting the guidelines unduly sacrifices performance of
the circuit. Therefore, the standard unit makes the following
exceptions.
Common-Mode EMI Filter Capacitors: The standard
supply uses 500 V capacitors to filter common-mode EMI.
NAVMAT guidelines would require 1000 V capacitors to meet
the 50% voltage derating (500 V dc input to output isolation),
resulting in less common-mode capacitance for the same space.
In typical electrical power supply systems, where the load
ground is eventually connected to the source ground, common-
mode voltages never get near the 500 V dc rating of the standard
supply. Therefore, a lower voltage rating capacitor (500 V)
was chosen to fit more capacitance in the same space in order
to better meet the conducted emissions requirement of MIL-
STD-461D (CE102). For those applications requiring 250 V
or less of isolation from input to output, the present designs
would meet NAVMAT guidelines.
Switching Transistors: 100 V MOSFETs are used in the
standard unit to switch the primary side of the transformers.
Their nominal off-state voltage meets the NAVMAT derating
guidelines. When the MOSFETs are turned off, however, mo-
mentary spikes occur that reach 100 V. The present generation
of MOSFETs are rated for repetitive avalanche, a condition that
was not considered by the NAVMAT deratings. In the worst
case condition, the energy dissipated during avalanche is 1% of
the device's rated repetitive avalanche energy. To meet the
NAVMAT derating, 200 V MOSFETs could be used. The
100 V MOSFETs are used instead for their lower on-state resis-
tance, resulting in higher efficiency for the power supply.
NAVMAT Junction Temperatures: The two types of power
deratings (current and temperature) can be independent of one
another. For instance, a switching diode can meet its derating
ADDC02828SA
REV. 0
12
ripple voltages will be smaller than if the load is grounded. The
test specifications do not state which procedure should be used.
However, in neither case (load grounded or floating) will the
typical EMI test setup described below be exactly representative
of the final system configuration EMI test. For the following
reasons, the same is true if separately packaged EMI filters are
used.
In almost all systems the output ground of the converter is ulti-
mately connected to the input ground of the system. The para-
sitic capacitances and inductances in this connection will affect
the common-mode voltage and the CE102 measurement. In
addition, the inductive impedance of this ground connection
can cause resonances, thereby affecting the performance of the
common-mode filter in the power supply.
In response to these ambiguities, the Analog Devices' converter
has been tested for CE102 under a constant load and with the
output ground floating. While these measurements are a good
indication of how the converter will operate in the final system
configuration, the user should confirm CE102 testing in the
final system configuration.
CE101: This test measures emissions on the input leads in the
frequency range between 30 Hz and 10 kHz. The intent of this
requirement is to ensure that the dc/dc converter does not cor-
rupt the power quality (allowable voltage distortion) on the
power buses present on the platform. There are several CE101
limit curves in MIL-STD-461D. The most stringent one app-
licable for the converter is that for submarine applications.
Figure 11 shows that the converter easily meets this requirement
(the return line measurement is similar). The components at
60 Hz and its harmonics are a result of ripple in the output of
the power source used to supply the converter.
CE102: This test measures emissions in the frequency range
between 10 kHz and 10 MHz. The measurements are made on
both of the input leads of the converter which are connected to
the power source through LISNs. The intent of this requirement
in the lower frequency portion of the requirement is to ensure
that the dc/dc converter does not corrupt the power quality
(allowable voltage distortion) on the power buses present on the
platform. At higher frequencies, the intent is to serve as a sepa-
rate control from RE102 on potential radiation from power
leads, which may couple into sensitive electronic equipment.
Figure 12 shows the CE102 limit and the measurement taken
from the +V
IN
line. While the measurement taken from the
input return line is slightly different, both comfortably meet the
MIL-STD-461D, CE102 limit. (Reference the last section of
EMI Considerations for how to adjust the external components
in the test setup circuit to increase the margin between the
specification limit and the measured results.)
CS101: This test measures the ability of the converter to reject
low frequency differential signals, 30 Hz to 50 kHz, injected on
the dc inputs. The measurement is taken on the output power
leads. The intent is to ensure that equipment performance is
not degraded from ripple voltages associated with allowable
distortion of power source voltage waveforms. Figure 7 shows a
typical audio susceptibility graph. Note that according to the
MIL-STD-461D test requirements, the injected signal between
30 Hz and 5 kHz has an amplitude of 2 V rms and from 5 kHz
to 50 kHz the amplitude decreases inversely with frequency to
0.2 V rms. The curve of the injected signal should be multiplied
by the audio susceptibility curve to determine the output ripple
of 70% of its maximum current, but its junction temperature
can be higher than 110
°
C if the case temperature of the con-
verter, which is not controlled by the manufacturer, is allowed
to go higher. Since some users may choose to operate the power
supply at a case temperature higher than 90
°
C, it then becomes
important to know the temperature rise of the hottest semicon-
ductors. This is covered in the specification table in the section
entitled Thermal Characteristics.
EMI CONSIDERATIONS
The ADDC02828SA has an integral differential- and common-
mode EMI filter that is designed to meet all applicable require-
ments in MIL-STD-461D when the power converter is installed
in a typical system setup (described below). The converter also
contains transient protection circuitry that permits the unit to
survive short, high voltage transients across its input power
leads. The purpose of this section is to describe the various
MIL-STD-461D tests and the converter's corresponding perfor-
mance. Consult factory for additional information.
The figures and tests referenced herein were obtained from
measurements on the ADDC02805SA, a single 5 V dc output
converter. Since the construction and topology of the 28 V dc
output converter is almost identical to the 5 V dc output converter,
and the component values of the EMI differential- and common-
mode filter in the 28 V dc output converter are identical to the
5 V output converter, the text references these figures and tests
as typical of the ADDC02828SA converter as well.
Electromagnetic interference (EMI) is governed by MIL-STD-461D,
which establishes design requirements, and MIL-STD-462D,
which defines test methods. EMI requirements are categorized
as follows (xxx designates a three digit number):
·
CExxx: Conducted Emissions (EMI produced internal to the
power supply, which is conducted externally through its input
power leads)
·
CSxxx: Conducted Susceptibility (EMI produced external to
the power supply, which is conducted internally through the
input power leads and may interfere with the supply's op-
eration)
·
RExxx: Radiated Emissions (EMI produced internal to the
power supply, which is radiated into the surrounding space)
·
RSxxx: Radiated Susceptibility (EMI produced external to
the power supply, which radiates into or through the power
supply and may interfere with its proper operation)
It should be noted that there are several areas of ambiguity, with
respect to CE102 measurements, that may concern the systems
engineer. One is the nature of the load. If it is constant, the
ripple voltage on the converter's input leads is due only to the
operation of the converter. If, on the other hand, the load is
changing over time, this variation causes an additional input
current and voltage ripple to be drawn at the same frequency. If
the frequency is high enough, the converter's filter will help
attenuate this second source of ripple, but if it is below approxi-
mately 100 kHz, it will not. The system may then not meet the
CE102 requirement, even though the converter is not the source
of the EMI. If this is the case, additional capacitance may be
needed across the load or across the input to the converter.
Another ambiguity in the CE102 measurement concerns common-
mode voltage. If the load is left unconnected from the ground
plane (even though the case is grounded), the common-mode
ADDC02828SA
REV. 0
13
at any frequency. When this is done, the worst case output
ripple at the frequency of the input ripple occurs at 5 kHz, at
which point there is typically a 25 mV peak-to-peak output
ripple.
It should be noted that MIL-STD-704 has a more relaxed require-
ment for rejection of low frequency differential signals injected on
the dc inputs than MIL-STD-461D. MIL-STD-704 calls for a
lower amplitude ripple to be injected on the input in a narrower
frequency band, 10 Hz to 20 kHz.
CS114: This test measures the ability of the converter to oper-
ate correctly during and after being subjected to currents in-
jected into bulk cables in the 10 kHz to 400 MHz range. Its
purpose is to simulate currents that would be developed in these
cables due to electromagnetic fields generated by antenna trans-
missions. The converter is designed to meet the requirements
of this test when the current is injected on the input power leads
cable. Consult factory for more information.
CS115: This test measures the ability of the converter to oper-
ate correctly during and after being subjected to 30 ns long
pulses of current injected into bulk cables. Its purpose is to
simulate transients caused by lightning or electromagnetic
pulses. The converter is designed to meet this requirement
when applied to its input power leads cable. Consult factory for
more information.
CS116: This test measures the ability of the converter to oper-
ate correctly during and after being subjected to damped sinu-
soid transients in the 10 kHz to 100 MHz range. Its purpose is
to simulate current and voltage waveforms that would occur
when natural resonances in the system are excited. The con-
verter is designed to meet this requirement when applied to its
input power leads cable. Consult factory for more information.
RE101: This requirement limits the strength of the magnetic
field created by the converter in order to avoid interference with
sensitive equipment located nearby. The measurement is made
from 30 Hz to 100 kHz. The most stringent requirement is for
the Navy. Figure 13 shows the test results when the pickup coil
is held 7 cm above the converter. As can be seen, the converter
easily meets this requirement.
RE102: This requirements limits the strength of the electric
field emissions from the power converter to protect sensitive
receivers from interference. The measurement is made from
10 kHz to 18 GHz with the antenna oriented in the vertical
plane. For the 30 MHz and above range, the standard calls for
the measurement to be made with the antenna oriented in the
horizontal plane as well.
In a typical power converter system setup, the radiated emis-
sions can come from two sources: 1) the input power leads as
they extend over the two meter distance between the LISNs and
the converter, as required for this test, and 2) the converter
output leads and load. The latter is likely to create significant
emissions if left uncovered since minimal EMI filtering is
provided at the converter's output. It is typical, however,
that the power supply and its load would be contained in a
conductive enclosure in applications where this test is applicable.
A metal screen enclosure was therefore used to cover the con-
verter and its load for this test.
Figure 14 shows test results for the vertical measurement and
compares them against the most stringent RE102 requirement;
the horizontal measurement (30 MHz and above) was similar.
As can be seen, the emissions just meet the standard in the
18 MHz28 MHz range. This component of the emissions is
due to common-mode currents flowing through the input power
leads. As mentioned in the section on CE102 above, the level
of common-mode current that flows is dependent on how the
load is connected. This measurement is therefore a good indi-
cation of how well the converter will perform in the final con-
figuration, but the user should confirm RE102 testing in the
final system. (Reference the last section of EMI Considerations
for how to adjust the external components in the test setup
circuit to increase the margin between the specification limit
and the measured results.)
RS101: This requirement is specialized and is intended to check
for sensitivity to low frequency magnetic fields in the 30 Hz to
50 kHz range. The converter is designed to meet this require-
ment. Consult factory for more information.
RS103: This test calls for correct operation during and after the
unit under test is subjected to radiated electric fields in the 10 kHz
to 40 GHz range. The intent is to simulate electromagnetic
fields generated by antenna transmissions. The converter is
designed to meet this requirement. Consult factory for more
information.
Circuit Setup for EMI Test
Figure 15 shows a schematic of the test setup used for the EMI
measurements discussed above. The output of the converter is
connected to a resistive load designed to draw full power. There
is a 0.1
µ
F capacitor placed across this resistor that typifies by-
pass capacitance normally used in this application. At the input
of the converter there are two differential capacitors (the larger
one having a series resistance) and two small common-mode
capacitors connected to case ground. The case itself was con-
nected to the metal ground plane in the test chamber. For the
RE102 test, a metal screen box was used to cover both the con-
verter and its load (but not the two meters of input power lead
cables). This box was also electrically connected to the metal
ground plane.
With regard to the components added to the input power lines,
the 100
µ
F capacitor with its 1
series resistance is required to
achieve system stability when the unit is powered through the
LISNs, as the MIL-STD-461D standard requires. These
LISNs have a series inductance of 50
µ
H at low frequencies,
giving a total differential inductance of 100
µ
H. As explained
earlier in the System Instability section, such a large series
source inductance will cause an instability as it interacts with the
converter's negative incremental input resistance unless some
corrective action is taken. The 100
µ
F capacitor and 1
resis-
tor provide the stabilization required.
It should be noted that the values of these stabilization components
are appropriate for a single converter load. If the system makes
use of several converters, the values of the components will need
to be changed slightly, but not such that they are repeated for
every converter. It should also be noted that most system appli-
cations will not have a source inductance as large as the 100
µ
H
built into the LISNs. For those systems, a much smaller input
capacitor could be used.
Increasing Margin Between Specification Limit and Measured
Results
With regard to the 2
µ
F differential-mode capacitor and the two
82 nF common-mode capacitors, these components were in-
cluded in the test setup to augment the performance of the
ADDC02828SA
REV. 0
14
power supply's internal EMI filter. The values were chosen to
achieve the results shown in Figures 12 and 14. To increase the
margin between the specification limits and the measured emis-
sions, larger external component values could be used.
To do this it is useful to know that most of the emissions below
10 MHz, whether conducted or radiated, are due to differential-
mode currents flowing in the input power leads. To make the
emissions in this frequency range smaller, the differential
capacitor value should be increased above 2
µ
F. Conversely,
most of the emissions above 10 MHz are due to common-mode
currents, and to make them smaller the common-mode capaci-
tors should be increased above the 82 nF value. In both cases it
is important to minimize the parasitic inductance of the capaci-
tors; the use of several smaller capacitors connected in parallel is
one way to achieve this.
Using larger valued capacitors than those shown in Figure 15 is
a good solution if an additional 6 dB10 dB of margin is de-
sired. However, if in an extremely sensitive application it is
desired to increase the margin by 20 dB or more, it may be
better to add both differential- and common-mode inductors to
the external components to make a higher order filter.
RELIABILITY CONSIDERATIONS
MTBF (Mean Time Between Failure) is a commonly used
reliability concept that applies to repairable items in which
failed elements are replaced upon failure. The expression for
MTBF is
MTBF = T/r
where
T = total operating time
r = number of failures
In lieu of actual field data, MTBF can be predicted per
MIL-HDBK-217.
MTBF, Failure Rate, and Probability of Failure: A proper
understanding of MTBF begins with its relationship to lambda
(
), which is the failure rate. If a constant failure rate is assumed,
then MTBF = 1/
, or
= 1/MTBF. If a power supply has an
MTBF of 1,000,000 hours, this does not mean it will last
1,000,000 hours before it fails. Instead, the MTBF describes
the failure rate. For 1,000,000 hours MTBF, the failure rate
during any hour is 1/1,000,000, or 0.0001%. Thus, a power
supply with an MTBF of 500,000 hours would have twice the
failure rate (0.0002%) of one with 1,000,000 hours.
What users should be interested in is the probability of a power
supply not failing prior to some time t. Given the assumption of
a constant failure rate, this probability is defined as
R(t )
=
e
t
where R(t) is the probability of a device not failing prior to some
time t.
If we substitute
= 1/MTBF in the above formula, then the
expression becomes
R(t )
=
e
t
MTBF
This formula is the correct way to interpret the meaning of
MTBF.
If we assume t = MTBF = 1,000,000 hours, then the probability
that a power supply will not fail prior to 1,000,000 hours of use
is e
1
, or 36.8%. This is quite different from saying the power
supply will last 1,000,000 hours before it fails. The probability
that the power supply will not fail prior to 50,000 hours of use is
e
.05
or 95%. For t = 10,000 hours the probability of no failure
is e
.01
or 99%.
Temperature and Environmental Factors: Although the
calculation of MTBF per MIL-HDBK-217 is a detailed process,
there are two key variables that give the manufacturer signi-
ficant leeway in predicting an MTBF rating. These two vari-
ables are temperature and environmental factor. Therefore, for
users to properly compare MTBF numbers from two different
manufacturers, the environmental factor and the temperature
must be identical. Contact the factory for MTBF calculations
for specific environmental factors and temperatures.
MECHANICAL CONSIDERATIONS
When mounting the converter into the next higher level assem-
bly, it is important to ensure good thermal contact is made
between the converter and the external heat sink. Poor thermal
connection can result in the converter shutting off, due to the
temperature shutdown feature (Pin 9), or reduced reliability for
the converter due to higher than anticipated junction and case
temperatures. For these reasons the mounting tab locations
were selected to ensure good thermal contact is made near the
hot spots of the converter, which are shown in the shaded areas
of Figure 24.
Figure 24. Hot Spots (Shaded Areas) of DC/DC Converter
The pins of the converter are typically connected to the next
higher level assembly by bending them at right angles, either
down or up, and cutting them shorter for insertion in printed
circuit board through holes. In order to maintain the hermetic
integrity of the seals around the pins, a fixture should be used
for bending the pins without stressing the pin-to-sidewall seals.
It is recommended that the minimum distance between the
package edge and the inside of the pin be 100 mils (2.54 mm)
for the 40 mil (1.02 mm) diameter pins; 120 mils (3.05 mm)
from the package edge to the center of the pin as shown in
Figure 25.
ADDC02828SA
REV. 0
15
DISABLE
ENB
GND
V
CC
OUT
HH73R-1MHZ
CONNER/WINFIELD
C1
.1µF
C2
100pF
+5V
SYNC OUT
SYNC 1
V
AUX
SYNC 2
SYNC 10
D2
1N5819
R13
1k
R5
1k
R2
1k
C3
.1µF
D1
1N966B
D3
1N5819
D4
1N5819
C4
100pF
C5
100pF
C13
100pF
R4
51k
OLH5801
(ISOLINK)
R3
357
SYNC IN
SYNC IN
2
3
5
6
8
R1
100k
A
K
A
K
Figure 28. Fault Tolerant, Secondary Side-Powered SYNC
Drive Circuit
NOTES
1. Input to Output Isolation: With the use of the Isolink opto-
coupler, we can use the output of the converters to power the
Conner/Winfield 1 MHz clock (output referenced) and the
Vaux pin (input referenced) to power the opto-coupler.
2. Fault Tolerant: All outputs are capacitively coupled to ensure
that if the master clock stops working the individual units will
continue to operate at their own internal clock frequency,
thereby eliminating a potential single point failure.
3. Radiated Emissions: C2 can be added to slow down the clock
edges (Tr and Tf) for reducing radiated emissions.
4. Table: The following table shows the capacitor and resis-
tor value to be used for the number of converters to be
synchronized.
Resistor
# of
Capacitor
Value
Converters
Value (pF)
(ohms)
1
1000
100
2
470
200
3
330
300
4
270
400
5
220
500
6
180
600
7
150
700
8
120
800
9
100
900
10
100
1K
0.100"
(2.54mm)
0.120"
(3.05mm)
Figure 25. Minimum Bend Radius of 40 Mil (1.02 mm) Pins
1
2
10
11
17
16
15
14
13
12
ADDC02828SA
PS1
C1
+28VDC
28RTN
RLOAD
NOTE: VALUE OF C1 IS DEPENDENT ON SOURCE IMPEDANCE.
REFER TO SECTION ON SYSTEM INSTABILITY CONSIDERATIONS.
Figure 26. Typical Power Connections and External Parts
for Converter (+28 V dc Out)
1
2
8
10
11
17
16
15
14
13
12
ADDC02828SA
1
2
8
10
11
17
16
15
14
13
12
ADDC02828SA
PS2
PS1
I
SHARE
C1
+28VDC
28RTN
+SENSE PS1
+SENSE PS2
RLOAD
+V
OUT
RETURN
NOTE: VALUE OF C1 IS DEPENDENT ON SOURCE IMPEDANCE.
REFER TO SECTION ON SYSTEM INSTABILITY CONSIDERATIONS.
Figure 27. Typical Connections for Paralleling Two
Converters (+28 V dc Out)
ADDC02828SA
REV. 0
16
C29947.52/97
PRINTED IN U.S.A.
Screening Levels for ADDC02828SA
Screening Steps
Industrial (KV)
Ruggedized Industrial (TV)
MIL-STD-883B/SMD (TV/883B)
Pre-Cap Visual
100%
MIL-STD-883, TM2017
Temp Cycle
N/A
N/A
Constant Acceleration
N/A
N/A
Fine Leak
Guaranteed to Meet
Guaranteed to Meet
MIL-STD-883, TM1014
MIL-STD-883, TM1014
Compliant to MIL-PRF-38534
Gross Leak
Guaranteed to Meet
Guaranteed to Meet
MIL-STD-883, TM1014
MIL-STD-883, TM1014
Burn-In
N/A
MIL-STD-883, TM1015,
96 Hrs at +115
°
C Case
Final Electrical Test
At +25
°
C, Per Specification
At +25
°
C, Per Specification
Table
Table
NOMINAL CASE DIMENSIONS IN INCHES AND (mm)
[All tolerances
±
.005" (
±
.13 mm) unless otherwise specified]
0.150 (3.81)
0.100 (2.54)
8 PLCS
0.200 (5.08)
0.150 (3.81)
0.200 (5.08)
0.390
±
0.010
(9.91
±
0.25)
0.800
±
0.010
(20.32
±
0.25)
1.145 (29.08)
2 PLCS
0.150 (3.81)
4 PLCS
TOP VIEW
0.149 (3.78)
DIA TYP
0.300 (7.62) SQ
±
0.010
4 PLCS
0.200 (5.08) 5 PLCS
0.250 (6.35)
2 PLCS
1.500
±
0.010
(38.10
±
0.25)
0.040
±
0.003
(1.02
±
0.08)
0.090
±
0.010
(2.29
±
0.25)
4 PLCS
2.745
±
0.010
(69.72
±
0.25)
1.800
(45.72)
TYP
2.100
±
0.010
(53.34
±
0.25)
5. All pins are a minimum length of 0.740 inches (18.80 mm)
when the product is shipped. The pins are typically bent up
or down and cut shorter for proper connection into the user's
system.
6. All pin-to-sidewall spacings are guaranteed for a minimum of
500 V dc breakdown at standard air pressure.
7. The case outline was originally designed using inch-pound units
of measurement. In the event of conflict between the metric
and inch-pound units, the inch-pound shall take precedence.
NOTES
1. The final product weight is 85 grams maximum.
2. The package base material is made of molybdenum and is
nominally 40 mils (1.02 mm) thick. The "runout" is less
than 2 mils per inch (0.02 mm per cm).
3. The high current pins (1017) are 40 mil (1.02 mm) diameter;
are 99.8% copper; and are plated with gold over nickel.
4. The signal carrying pins (19) are 18 mil (0.46 mm) diam-
eter; are Kovar; and are plated with gold over nickel.
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